Rapid progress in telecommunication and radar technology is placing increasing demands on wireless system performance. Such demands include high information capacity and the ability to track multiple mobile targets or users.1–3 Military research programs in particular are placing heavy emphasis on Network Centric Warfare, a wireless network concept in which numerous theater assets exchange vast amounts of data using wideband electronically scanned phased arrays. A parallel military research thrust seeks to reduce radar cross-section and terminal crowding by developing multi-function and multi-frequency phased arrays.4–7

To meet these new challenges, this article introduces a T/R module for multi-frequency phased array applications. The module is capable of operation at both X- and Ku-band with four frequency channels. Two channels at 10 and 19 GHz are for the transmitter and another two at 12 and 21 GHz are for the receiver. All four channels are full duplex to allow simultaneous transmission and reception. These operating frequencies are chosen to demonstrate the nature of the technology; other frequencies can be selected by adjusting the design according to a systematic procedure. A key feature of the T/R module is a broadband microstrip multiplexer that duplexes the transmit and receive signals. The multiplexer directs the transmit signals through a broadband MMIC power amplifier (PA) and the receive signals through a broadband MMIC low noise amplifier (LNA). The multiplexer design uses an innovative yet straightforward technique to efficiently suppress the parasitic passbands of the 10 and 12 GHz channels that would otherwise interfere with the channels at 19 and 20 GHz. Measured results indicate that the module’s total output power at 1 dB compression is 21 dBm with a gain of 16.9 dB at 10 GHz and 15.1 dBm with a gain of 17.2 dB at 19 GHz. The noise figure is 6.4 dB with a gain of 23.9 dB at 12 GHz and 10.9 dB with a gain of 17.6 dB at 21 GHz. Broadband MMIC technology is advancing rapidly, and these results can be improved considerably by taking advantage of several recent advances in MMIC technology that are noted in this article. The module is intended for use in multi-frequency/multi-function phased array systems. The authors expect that the design approach detailed in this article will stimulate further advances in microwave front-end designs for these applications.

Module Design

Figure 1 presents the layout of the four-channel T/R module. Two identical microstrip multiplexers fabricated on RT-Duroid 5880 (?r = 2.2) are used to route the transmit and receive signals through the MMIC-based amplifier assemblies. The receive signals at 12 and 21 GHz pass through ports 1 and 3 of Multiplexer A and are amplified by a broadband LNA before passing through ports 3 and 1 of Multiplexer B. Transmit signals at 10 and 19 GHz pass from an up-converter through ports 1 and 2 of Multiplexer B and are amplified by a broadband PA. This section describes the design approach behind the multiplexer and the integration of the MMICs within the module.

Fig. 1 Multi-channel T/R module layout.


Within each multiplexer are four parallel-coupled microstrip bandpass filters with different passband frequencies. Each channel’s passband frequency can be changed by modifying the lengths of the resonators within the relevant bandpass filter, while adjusting the phase balances among all of the filters and T-junctions. As stated previously, the transmit frequencies are chosen as 10 and 19 GHz, and the receive frequencies as 12 and 21 GHz. Simulations, however, show that this topology works well for channel frequencies chosen across the 1 through 28 GHz range.

The multiplexer needs to provide high isolation among all four channels in order to allow simultaneous transmit/receive operation. In fact, to avoid unintended feedback oscillations, both multiplexers must together provide sufficient isolation across all frequencies to suppress the gain at any particular frequency around the T/R loop. To do this, it is critical that the multiplexer efficiently suppress the second-harmonic parasitic passbands of the 10 and 12 GHz filters. Several schemes for improving the second-harmonic suppression of microstrip bandpass filters have recently been proposed, including “wiggly-line” designs.8 This design adopts a straightforward alternative,9,10 as illustrated in Figure 2.

Fig. 2 Harmonic-suppressed microstrip parallel-coupled banpass filter.

Typically, the performance of a microstrip bandpass filter depends on the number of resonators and on the dimensions W, S, and L for each section of the coupled resonators. For a given substrate, the passband center frequency decreases as the resonator length L increases, and the passband bandwidth decreases with increasing W and S, as the resonator Q increases. For a given resonator length L, the parasitic passband can be controlled by modifying the dimensions W and S together with the even/odd impedances of the input and output coupled resonators. In practice, this is easily done, beginning with a standard microstrip bandpass filter design. The design can then be tuned with a planar electromagnetic simulator by directly adjusting the dimensions W and S of the individual resonators in order to increase the harmonic suppression of the filter at twice the passband frequency. Simulation and experiment indicate that the second-harmonic rejection is improved primarily by decreasing the width W of the input/output coupled resonators to less than the width of the input/output microstrip line while maintaining strong coupling to the input/output resonators by reducing the gap S. Further details on the approach can be found in the literature.9,10 As illustrated in Figure 3, the second-harmonic rejection of a microstrip bandpass filter, etched on RT-Duroid 5880, can be increased by 28 dB at 21 GHz with a corresponding increase in insertion loss of only 0.7 dB. The dimensions of the harmonic-suppressed filter are given in Table 1.

Fig. 3 Comparison between a general microstrip banpass filter and a second harmonic-suppressed microstrip banpass filter.

The complete multiplexer contains four second-harmonic-suppressed bandpass filters connected together. The designed dimensions of each of the individual filters are given in Table 2. For the overall multiplexer, it is necessary to carefully design the connecting lines and tee junctions in order to maintain the proper phase relationships at all four channel frequencies. This can be done in any standard microwave CAD package by using microstrip line elements to connect together the S-parameter files for the individually designed microstrip filters. The lengths of the microstrip lines are adjusted using the optimization engine in order to balance the phase for each frequency channel at the correct multiplexer ports. The final design is then simulated in Zeland IE3D, a planar electromagnetic simulator,11 and measured on an Agilent 8510C network analyzer. Figure 4 confirms that measurement and calculation compare favorably. The insertion losses at 10, 12, 19 and 21 GHz are 1.81, 1.90, 2.88 and 2.51 dB, respectively, and the return loss is 20.6 dB at 10 GHz, 10.7 dB at 12 GHz, 22 dB at 19 GHz and 16.5 dB at 21 GHz. The measured isolation among all channels is greater than 32 dB. The isolation achieved is at least 15 dB higher than can be achieved using a typical circulator-based duplexer across this same frequency range.

Fig. 4 Measured insertion loss of the four-channel microstrip multiplexer.

MMIC Amplifiers

The T/R module uses the two multiplexers to separate the transmit and receive signals for amplification by broadband MMIC-based gain blocks. The integration of the MMIC amplifiers within the T/R module is illustrated in Figure 5. Each MMIC amplifier is solder-mounted on an individual Cu-Mo carrier to allow for individual testing as needed. Alumina thin-film networks (TFN) provide connection pads for the bias wires and 50 ? lines to the RF inputs and outputs on the MMICs. A 10-mil gold ribbon is used for all RF connections between the MMICs, TFNs and multiplexers. External bias capacitors are mounted onto the carriers with conductive epoxy and bonded to the MMICs, with care taken to minimize the bond wire lengths in order to ensure low frequency stability.

Fig. 5 Intergration of the MMIC-based amplifiers within the T/R module.

The MMIC amplifiers used in this study are standard products available from TriQuint Semiconductor. The transmit channel amplifiers are chosen to provide moderate amplification with high P1dB specifications across the band, while the receive channel amplifiers are chosen to provide high gain with moderate noise figure across the band.

In the transmit path, the TGA8310 acts as an initial gain stage, feeding a TGA8334 that has enough P1dB to drive the final TGA8334 to saturation. The calculated overall gain for the three-stage power amplifier is 23 dB at both 10 and 19 GHz. The P1dB of the PA block is determined by the compression of an end-stage chip amplifier (TGA8334). In the receive path, two TGA1342s feed a TGA8310. The calculated gain of the three-stage low noise amplifier (LNA) is 27 dB at 12 GHz and 24 dB at 21 GHz. The noise figure of a multi-stage amplifier is given by12

where Fn, Mn and Gpn are the noise figure, input impedance mismatch and power gain, respectively, of an n-stage amplifier. Substituting the typical noise figures, mismatch and gain values for Fn, Mn and Gpn into Equation 1, the calculated overall noise figure of the three-stage low noise amplifier is 4.2 dB at 12 GHz and 8.3 dB at 21 GHz.

Figure 6 shows the measured gains of the broadband LNA and PA. The measured gain of the three-stage PA is 20.5 dB at 10 GHz and 22.6 dB at 19 GHz. The measured output power P1dB is 23 dBm at 10 GHz and 18.5 dBm at 19 GHz. The measured gain of the three-stage low noise amplifier is 27.5 and 23.5 dB at 12 and 21 GHz, respectively. The measured noise figure of the amplifier is 4.4 dB at 12 GHz and 8 dB at 21 GHz. These measurements agree very well with the above calculations.

Fig. 6 Measured gains of the broadband LNA and PA.

It is important to note that the performance of these TriQuint chips is far surpassed by new replacement products released by the same company in 2004. The new TGA2509 power amplifier provides 1 W of output power with 17 dB of gain across the 2 to 22 GHz band. Likewise, the new TGA2513 low noise amplifier provides 17 dB gain across the same band with a noise figure less than 2 dB at mid-band. The same integration approach used in this article can be applied to these new chips.

Module Performance

The small-signal response of the complete T/R module is shown in Figure 7. It indicates that the measured gain of the receive channels is 23.9 dB at 12 GHz and 17.6 dB at 21 GHz, the return loss is 12.4 dB at 12 GHz and 14 dB at 21 GHz, and the rejection at the transmit channels is 61 dB at 10.04 GHz and 32 dB at 19.14 GHz. It also shows that the gain and return loss of the transmit channels are 16.9 and 25 dB, respectively, at 10 GHz and 17.2 and 11.6 dB at 19 GHz. The rejection at the receive channel is 62 dB at 12 GHz and 40 dB at 21 GHz. The measured output power P1dB, shown in the power compression curves given in Figure 8, is 21 dBm at 10 GHz and 15.1 dBm at 19 GHz. The measured noise figure of the T/R module is 6.4 dB at 12 GHz and 10.9 dB at 21 GHz. A comparison of calculated and measured results for the T/R module is listed in Table 3. The measured and calculated results agree well.

Fig. 7 Small-signal response of the T/R module.

Fig. 8 Measured PldB of the transmitter portion of the T/R module.

The intended application of the T/R module is in a multi-frequency phased array transceiver. One of the most important system requirements in phased array applications is the amplitude and phase balance among the T/R modules. Excellent amplitude and phase balance play an important role in the system performance, allowing the phased array to achieve a low side-lobe level, optimal system gain and good beam scanning. The measured gains and phases of four T/R modules manufactured for this project are shown in Table 4. The maximum absolute variation in gain amplitude and phase among the four T/R modules is 2.7 dB and 38.7° at 10 GHz, 1 dB and 38° at 12 GHz, 1.1 dB and 31° at 19 GHz and 1.6 dB and 39° at 21 GHz. The maximum absolute variation in gain amplitude among the T/R modules is less than 0.5 dB when gain control circuits are adopted. The phased array was measured using an HP 85301A antenna measurement system in an anechoic chamber. Results indicate that the beam scanning of the phased array is over ±27° at each of the four channels.5


A broadband multi-frequency T/R module with four channels at 10, 12, 19 and 21 GHz is presented in this article. The T/R module consists of two multi-frequency harmonic-suppressed microstrip multiplexers, a broadband MMIC low noise amplifier and a broadband MMIC power amplifier. Measurements show that the T/R module demonstrates excellent performance over a wide frequency range from 10 through 21 GHz. The passband frequencies of the four channels of the T/R module can be easily changed within the 2 through 21 GHz range by simply adjusting the passband frequencies of the multiplexers. Application in a broadband and multi-frequency phased array further demonstrates that the T/R module works very well. This broadband multi-channel approach should have many applications in emerging military communications systems.


The authors wish to thank Ming-yi Li and Tae-Yeoul Yun of Texas A&M University for their assistance in the phased array application; James Klein and Brad Heimer at Raytheon for donating the assembly of the MMIC carriers; and James Carroll of TriQuint for donating the MMICs. This work was supported in part by the US Air Force and RST Scientific Research Inc.


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Chunlei Wang received his BS and MS degrees from the University of Electronic Science and Technology of China, Chengdu, China, in 1982 and 1990, respectively. He was honored by the National Electronic Science and Technology Progress Awards in China in 1984, 1987 and 1995. In 1992, he was honored as the “Outstanding Youth of China Electronic Industry” by the Ministry of Electronic Industry of China. He has been a senior engineer with WiseWave Technologies Inc., Torrance, CA, since 2003. His current interest includes the design of transceivers operating up to 86 GHz for last mile wireless access and large-capacity communications with data rates greater than 1 Gbts. He has published more than 30 papers and is a senior member of the IEEE.

Christopher T. Rodenbeck received his BS, MS and PhD degrees in electrical engineering from Texas A&M University, College Station, TX, in 1999, 2001 and 2004, respectively. He is currently a senior member of the technical staff at Sandia National Laboratories, Albuquerque, NM, where he works on the design of microwave circuits, antennas and phased arrays. During the summer months of 1998, 1999 and 2000, he was with TriQuint Semiconductor, Dallas, TX, as an intern in the MMIC design group. He has authored or co-authored over 25 papers in the microwave and millimeter-wave field.

Kai Chang received his BSEE degree from National Taiwan University, Taipei, Taiwan, his MS degree from the State University of New York at Stony Brook and his PhD degree from the University of Michigan, Ann Arbor, MI, in 1970, 1972 and 1976, respectively. He joined the electrical engineering department of Texas A&M University in August 1985 as an associate professor and was promoted to professor three years later. In January 1990, he was appointed e-systems endowed professor of electrical engineering. His current interests include microwave and millimeter-wave devices and circuits, microwave integrated circuits, integrated antennas, wideband and active antennas, phased arrays, microwave power transmission and microwave optical interactions. He has authored or co-authored several books and is the editor of Microwave and Optical Technology Letters and the Wiley Book Series in Microwave and Optical Engineering (over 65 books published). He has published over 400 papers, and several book chapters in the areas of microwave and millimeter-wave devices, circuits and antennas.

Matthew R. Coutant received his BS degree in electrical engineering from Oklahoma State University, Stillwater, OK, and his MS degree from Texas A&M University, College Station, TX, in 1997 and 2000, respectively. He is currently working toward his PhD degree at Texas A&M, while also working as a MMIC designer at TriQuint Semiconductor in Dallas, TX.