Microwave Journal

200 W High Efficiency 1.8 to 2.7 GHz GaN HEMT Doherty Amplifiers for Cellular

November 14, 2018

In next-generation 4G/5G telecommunication systems, it is required that the radio transmitter and the power amplifier-the key part of the transmitter-operate with high efficiency over a wide frequency range, to provide multi-band and multi-standard concurrent operation. In these systems, with increased bandwidths and high data rates, the transmitting signal is characterized by high peak-to-average power ratio (PAPR) due to wide and rapid variations of the instantaneous transmitting power. It is very important to provide high efficiency at both maximum output power and lower power levels, typically ranging from 6 dB back-off and less, over a wide frequency bandwidth. Different 3GPP LTE-Advanced bands for 4G/5G systems, with up to 40 MHz channel bandwidths, are expected to be covered: tri-band (SMH, CLR, GSM) from 0.7 to 1 GHz, tri-band (DCS, PCS, IMT) from 1.8 to 2.2 GHz, dual-band (IMT and IMT-e) from 2.1 to 2.7 GHz or multi-band from 1.8 to 2.7 GHz. By using GaN HEMT technology and innovative Doherty architectures, average efficiencies of 50 to 60 percent for output powers up to 200 W can be achieved, significantly reducing transmitter cost, size and power consumption.

For a conventional Doherty amplifier with a quarter-wave impedance transformer and a quarter-wave output combiner, the measured power-added efficiency of 31 percent at about 43 dBm-6 to 7 dB back-off from the saturated output power-has been achieved across the frequency range from 1.5 to 2.14 GHz.1 To improve the broadband performance of the conventional Doherty amplifier, the output network can be composed of two quarter-wave impedance inverters with reduced impedance transformation ratios.2 For broadband combining, an output quarter-wave transmission line with fixed characteristic impedance can be replaced by a multi-section transmission line with different characteristic impedances and electrical lengths, which enables frequency coverage from 2.2 to 2.96 GHz.3 In this case, broadband matching is realized by applying the simplified real frequency technique with the desired frequency-dependent optimum impedances. However, nonlinear optimization of the entire Doherty amplifier system makes the design complicated to simulate and results in a large size for the final board implementation.

A high peak power of 350 W has been achieved across the lower frequency band of 760 to 960 MHz using a modified combining scheme with two quarter-wave lines in the peaking path.4 Using an asymmetric Doherty architecture, saturated power greater than 270 W, linear gain greater than 13 dB and a drain efficiency greater than 45 percent at 8 dB back-off has been achieved across 2.5 to 2.7 GHz.5


A multi-band Doherty amplifier can be achieved when all of its components are designed to provide their corresponding characteristics over the required bandwidth of operation. The carrier and peaking amplifiers should provide broadband, high efficiency performance when, for example, their input matching circuits are designed as broadband. The load network generally comprises a lowpass, lumped or transmission line structure with two or three matching sections. Therefore, it is very important for matching circuits to be partly implemented inside the device package to achieve an average output power of 40 W and higher, especially for the input matching circuit, given the very low device impedance across the required frequency range.

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Figure 1 Fig. 1 Equivalent circuit of the packaged device (a) and |S11| performance (b).

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Figure 2 Class AB PA with broadband conjugate matching (a) and measured performance (b).

Figure 1 shows the equivalent circuit of a device with input matching elements inside the package and the small-signal |S11| at the input of the internal input matching circuit, including the package lead frame. The 50 V device, fabricated by Sumitomo Electric Device Innovations, has six basic 15 W GaN HEMT cells connected in parallel and capable of providing a combined saturated output power of more than 80 W across the entire band from 1.8 to 2.7 GHz. The three-section microstrip transformer is implemented using an alumina substrate with a high permittivity of 250 and a thickness of 0.16 mm, yielding a compact structure transforming the device input impedance to 10 Ω, with an |S11| less than −25 dB.

Generally, the multi-band impedance transformer required for broadband operation is represented as a configuration of N cascaded transmission lines (N ≥ 2) with different characteristic impedances.6 For example, to match the output impedance of 25 Ω to a load impedance of 50 Ω, the broadband output transformer can be realized using a two-section microstrip line, where the characteristic impedance of the first quarter-wave section is 30 Ω, and the characteristic impedance of the second quarter-wave section is 42 Ω. In this case, the input impedance magnitude variation of ±0.5 Ω and phase variation of ±1 degree can be achieved from 2 to 2.8 GHz, simultaneously covering the 2.1 GHz (2.11 to 2.17 GHz) and 2.6 GHz (2.62 to 2.69 GHz) WCDMA/LTE bands.7 At the same time, magnitude variations of ±1 Ω and phase variations of ±2 degrees can be achieved over a 1 GHz bandwidth from 1.9 to 2.9 GHz, which means that reducing the mid-band frequency to 2.3 GHz can result in simultaneous tri-band operation, i.e., including an additional 1.8 GHz DCS/WCDMA/LTE band (1805 to 1880 MHz).

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Figure 3 Doherty PA using dual-path packaged transistor (a) and measured performance (b).

Figure 2a shows the simplified schematic of a single-ended, 80 W GaN HEMT power amplifier operating in class AB mode, with external input and output matching circuits, which operates from 1.7 to 2.7 GHz. The input and output matching circuits, implemented on an RO4350 substrate, represent a two-stepped microstrip line transformer, with each line section a different characteristic impedance and electrical length. The matching network provides a conjugate-match with the device input and equivalent output impedance at the fundamental frequency. With this design, an output power at 1 dB gain compression (P1dB) of greater than 48 dBm, a power gain greater than 12 dB and drain efficiency greater than 52 percent were measured across 1.8 to 2.7 GHz, as shown in Figure 2b. Previously, drain efficiencies greater than 60 percent were reported between 1.9 and 2.9 GHz with commercial 45 W GaN HEMT transistors, using the simplified real frequency technique to determine optimum impedances and element values for the highest efficiencies across the frequency range.8

The classic two-stage Doherty amplifier has limited bandwidth in the low-power region, since it is necessary to provide an impedance transformation from 25 to 100 Ω when the peaking amplifier is turned off. This results in a loaded quality factor QL = √100/25 − 1 = 1.73 at 3 dB output power reduction, which is sufficiently high for broadband operation. At high-power levels, achieving broadband output matching of the carrier and peaking amplifiers with a broadband output quarter-wave transformer, it is possible to maximize the frequency bandwidth.

Figure 4

Figure 4 Broadband inverted Doherty PA.

Figure 3a shows the circuit diagram of a conventional two-stage Doherty amplifier implemented on 20 mil RO4350 substrate and using two, 80 W GaN HEMT power transistors with internal input matching and metal-ceramic flange packages. The input and output matching circuits are microstrip lines of different electrical lengths and characteristic impedances for the two-stepped structures. The input splitter is a broadband coupled-line coupler from Anaren (model X3C17A1-03WS), which provides a maximum phase balance of ±5 degrees and an amplitude balance of ±0.5 dB from 690 to 2700 MHz. Figure 3b shows the measured power gain and drain efficiency at five in-band frequencies. A power gain of more than 9 dB is achieved from 1.8 to 2.7 GHz, with drain efficiencies of about 60 percent at an output power corresponding to 3 dB gain compression (except at the high end of the band) and between 40 and 50 percent at 6 dB back-off. Given the bandwidth limitations of the conventional structure, the Doherty effect is not strong across the full band, with lower efficiency at the higher frequencies.


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Figure 5 Load network (a) and impedance transformation (b).

Figure 4 shows the schematic of an inverted broadband Doherty amplifier with an impedance transformer using a quarter-wave line connected to the output of the peaking amplifier. This architecture can be helpful if, in the low-power region and depending upon the characteristics of the transistor, it is easier to provide a short circuit rather than an open circuit at the output of the peaking amplifier.9 In this case, a quarter-wave line is used to transform a very low output impedance after the offset line to a high impedance seen from the load junction. Taking into account the device package parasitics of the peaking amplifier, an optimized output matching circuit and a proper offset line can be designed to maximize the output power from the peaking device in the high-power region and approximate a short circuit in the low-power region.10

To better understand the operating principle of an inverted Doherty amplifier, consider the load network (see Figure 5a) when the peaking amplifier is turned off. In the low-power region, the phase adjustment of the offset line with electrical length θ causes the peaking amplifier to be short-circuited (ideally 0 Ω), and the matching circuit with offset line provides the required impedance transformation from 25 Ω to the high output impedance Zout seen by the carrier device output at 6 dB power back-off (ideally 100 Ω with the quarter-wave transformer), as shown in Figure 5b. In this case, the short circuit at the end of the quarter-wave line transforms to an open circuit at its input, preventing power leakage to the peaking path when the peaking transistor is turned off. In the high-power region, both carrier and peaking amplifiers are operating in parallel in a 50 Ω environment, and the output quarter-wave line with a characteristic impedance of 35.3 Ω transforms 25 Ω to the 50 Ω load.

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Figure 6 ZMatch (a) and ZPeaking (b) of the peaking amplifier from 1.8 to 2.7 GHz.

Using this configuration and two commercial 10 W GaN HEMT power transistors, the broadband inverted GaN HEMT Doherty amplifier was designed to achieve an average drain efficiency of 47 percent, average output power of 38 dBm (saturated power of 44 dBm) and a power gain of more than 11 dB across 1.8 to 2.7 GHz.7,11 The impedance at different points of the load network of the peaking amplifier when it is off are shown in Figure 6, where Zmatch (see Figure 6a) indicates low reactance at the output of the load network over the frequency range from 1.8 to 2.7 GHz, having near zero reactance at the mid-band frequency and some inductive and capacitive reactances when the operating frequency approaches the edges. By using a quarter-wave series transmission line, at higher frequencies an open circuit condition is provided with sufficiently high inductive and capacitive reactances across the band, indicated by Zpeaking in Figure 6b. Hence, the broadband performance of the inverted Doherty structure can be achieved in a practical realization.

Figure 7a shows the load network equivalent circuit for the carrier amplifier with the impedance Zcarrier seen by the carrier device, whose real component varies slightly around 10 Ω (see Figure 7b). Taking into account the device output shunt capacitance Cout of about 5 pF and series output inductance Lout provided by the bond wire and package lead frame inductances, the impedance seen by the device multi-harmonic current source across the frequency bandwidth of 1.8 to 2.7 GHz is increased by 2x from 5 Ω at the input of the broadband output impedance transformer, which is large enough to achieve high efficiency at back-off output power levels. In this case, the device output capacitance and bond wire inductor constitute a lowpass L-type matching section, increasing the load impedance seen internally by the device multi-harmonic current source at the fundamental frequency.

Figure 8 shows simulation results for the small-signal |S11| and |S21| versus frequency, demonstrating the bandwidth capability of a modified inverted transmission line GaN HEMT Doherty amplifier, covering 1.6 to 3 GHz with a power gain greater than 11 dB.

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Figure 7 Matching network (a) and load impedance (b) for the carrier amplifier.

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Figure 8 Simulated small-signal S-parameters vs. frequency.



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Figure 9 Simulated power gain and drain efficiency of the broadband inverted Doherty PA.

Figure 9 shows the simulated large-signal power gain and drain efficiency of a transmission line GaN HEMT tri-band inverted Doherty amplifier using a 20 mil RO4350 substrate, with the carrier gate bias Vgc = −2.5 V, peaking gate bias Vgp = −5.5 V and DC supply voltage VDD = 50 V. The design has an output power greater than 53 dBm and a linear power gain greater than 10 dB across the entire 1.8 to 2.7 GHz range. Drain efficiencies greater than 50 percent at saturation and 7 dB back-off are simulated at the center of the three bands-at 1.85, 2.15 and 2.65 GHz-with maximum drain efficiency greater than 70 percent at the lower frequencies and peak efficiency at maximum back-off output power of around 6 dB over the entire frequency range.

A test board of the tri-band inverted Doherty amplifier using two 80 W GaN HEMT power transistors with internal input matching and in metal-ceramic flange packages was fabricated on a 20 mil RO4350 substrate. The input splitter, a broadband Anaren model X3C17A1-03WS 90 degree hybrid coupler, has a maximum phase balance of ±5 degrees and amplitude balance of ±0.5 dB from 690 to 2700 MHz. The input matching circuit, output load network and gate and drain bias circuits (having bypass capacitors on their ends) were microstrip lines of different electrical lengths and characteristic impedances. The output lead inductances of the packaged GaN HEMT device were minimized.

table 1
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Figure 10 Measured performance of the broadband inverted Doherty PA.

Figure 10 shows the measured power gain and drain efficiency of the transmission line GaN HEMT inverted Doherty amplifier at five frequencies. A power gain greater than 9 dB was achieved from 1.8 to 2.7 GHz. Drain efficiencies greater than 55 percent at saturation (P3dB) and around 50 percent at 7 dB back-off were measured across the entire band, with the maximum drain efficiency greater than 70 percent at the frequencies below 1.95 GHz and peak efficiency points at maximum back-off power around 6 dB over the entire frequency range. The test conditions for concurrent transmission of a four carrier GSM signal and a 10 MHz LTE signal with a PAPR of 8 dB are shown in Table 1. A drain efficiency of 51 percent with an average total output power of 45.5 dBm (18.2 W for the GSM signal and 17 W for the LTE signal) were achieved using in-house digital predistortion (DPD) linearization. With dual-band DPD and the four carrier GSM signal, the out-of-band intermodulation levels were lower than −70 dBc. With the 10 MHz LTE signal, the adjacent channel leakage ratio (ACLR) was better than −57 dBc (see Figure 11).

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Figure 11 DPD linearization of the inverted Doherty amplifier with a 10 MHz LTE signal.


To provide multi-band and multi-standard operation, LTE and 5G base stations are requiring that a power amplifier operate with high efficiencies over increasingly wide frequency ranges. Using GaN HEMT transistors and innovative Doherty architectures, tri-band coverage (1.8 to 2.7 GHz) with powers to 200 W and average efficiencies of 50 to 60 percent can be achieved, significantly reducing transmitter cost, size and power consumption.


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